Wide Range Frequency Synthesizer with Quadrature Generation and Spur Cancellation

ABSTRACT

A frequency synthesizer generates a wide range of frequencies from a single oscillator while achieving good noise performance. A cascaded phase-locked loop (PLL) circuit includes a first PLL circuit with an LC voltage controlled oscillator (VCO) and a second PLL circuit with a ring VCO. A feedforward path from the first PLL circuit to the second PLL circuit provides means and signal path for cancellation of phase noise, thereby reducing or eliminating spur and quantization effects. The frequency synthesizer can directly generate in-phase and quadrature phase output signals. A split-tuned ring-based VCO is controlled via a phase error detection loop to reduce or eliminate phase error between the quadrature signals.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional Application No.61/622,977 entitled “Wide Range Frequency Synthesizer with QuadratureGeneration and Spur Cancellation” to Masum Hossain, et al., filed onApr. 11, 2012, the contents of which is incorporated by referenceherein.

BACKGROUND

Frequency synthesizers are commonly used in wireless communicationsystems for generating a range of frequencies from a single oscillator.In recent years, the number of different wireless bands and standards inwhich a mobile device may communicate has increased dramatically. Forexample, mobile devices may communicate using different standards suchas GSM/EDGE, 3G, 4G, WiFi, GPS, Bluetooth, and others, each of whichutilize different frequency bands. However, traditional frequencysynthesizers are unable to generate low noise signals over a large rangeof frequencies. While some mobile devices overcome this problem byutilizing multiple frequency synthesizers tuned to different frequencyranges, this traditional design comes at a substantial area and powerpenalty.

BRIEF DESCRIPTION OF THE DRAWINGS

The teachings of the embodiments herein can be readily understood byconsidering the following detailed description in conjunction with theaccompanying drawings.

FIG. 1 illustrates a frequency synthesizer having a cascaded PLLarchitecture according to one embodiment.

FIG. 2 illustrates a PLL circuit according to one embodiment.

FIG. 3 illustrates waveforms associated with operation of a PLL circuitaccording to one embodiment.

FIG. 4 illustrates a frequency synthesizer circuit having a cascaded PLLarchitecture according to one embodiment.

FIG. 5 illustrates a phase-domain linear representation of a cascadedPLL circuit according to one embodiment.

FIG. 6A illustrates a PLL circuit with a split-tuned VCO according toone embodiment.

FIG. 6B illustrates a split-tuned LC-based VCO according to oneembodiment.

FIG. 7 illustrates a phase error correction module for a PLL circuitaccording to one embodiment.

FIG. 8A illustrates a process for frequency synthesis using a cascadedPLL circuit with a feedforward path according to one embodiment.

FIG. 8B illustrates a process for frequency synthesis using a cascadedPLL circuit with an LC-based PLL and a ring-based PLL according to oneembodiment.

FIG. 8C illustrates a process for frequency synthesis using a PLLcircuit producing two output signals having a relative phase shiftaccording to one embodiment.

FIG. 8D illustrates a process for frequency synthesis using a PLL havinga phase interpolator according to one embodiment.

DETAILED DESCRIPTION OF EMBODIMENTS

A frequency synthesizer generates a wide range of frequencies from asingle oscillator while achieving good noise performance. The frequencysynthesizer can therefore be used in a multiband wireless transceivercompatible with a variety of different wireless standards. In oneembodiment, for example, a frequency range of 900 MHz to 6 GHz isachieved to enable compatibility with standards such as, GSM/GPRS/EDGE,WCDMA, RFID, ZigBee, UWB, 802.11 and 802.16.

In one embodiment, the frequency synthesizer comprises a cascadedphase-locked loop (PLL) circuit in which a first PLL circuit has an LCvoltage controlled oscillator (VCO) and a second PLL circuit has a ringVCO. A feedforward path from the first PLL circuit to the second PLLcircuit improves phase noise, thereby reducing or eliminating spur andquantization effects. Furthermore, an embodiment of the frequencysynthesizer directly generates in-phase and quadrature phase outputsignals and uses a split-tuned ring-based VCO to reduce or eliminate I-Qphase error. Various embodiments of the frequency synthesizer aredescribed in further detail below.

Cascaded Phase-Locked Loop Circuit

FIG. 1 illustrates a high level block diagram of a cascaded phase-lockedloop (PLL) architecture 100 for a frequency synthesizer. Cascaded PLLarchitecture 100 comprises a first PLL circuit 110, a second PLL circuit120, and a feedforward control circuit 130. First PLL circuit 110receives a reference signal 102 having a reference frequency f_(REF),and generates a control signal 112 and an intermediate signal 104 havinga second frequency f_(INT). Second PLL circuit 120 receives intermediatesignal 104 having the second frequency f/NT and receives a feedforwardsignal 114. Second PLL circuit 120 generates an output signal 106 havinga frequency f_(OUT), based on feedforward signal 114 and intermediatesignal 104. Furthermore, feedforward control circuit 130 receivescontrol signal 112 from first PLL circuit 110 and generates feedforwardsignal 114. In one embodiment, control signal 112 represents noisegenerated by first PLL 110 (e.g., due to quantization error). Based onthis control signal 112, feedforward control circuit 130 generatesfeedforward signal 114 that adjusts a parameter of second PLL 120 suchthat the noise is canceled or reduced in output signal 106.

In one embodiment, first PLL circuit 110 comprises a fractional-NLC-based PLL circuit such as, for example, the PLL circuit 200 of FIG. 2described in further detail below. Furthermore, in one embodiment,second PLL circuit 120 comprises an integer-N ring-based PLL circuitsuch as, for example, the PLL circuit 420 of FIG. 4 or the circuits ofFIG. 6 or FIG. 7 described in further detail below. Overall, thecascaded PLL circuit 100 in this configuration functions as afractional-N PLL because fractional control of the first PLL 110 enablesfractional frequencies to be achieved at the output 106. This particularconfiguration is furthermore advantageous for achieving a frequencysynthesizer with both a wide frequency range and good phase noise.Generally, LC-based PLLs have relatively limited tuning ranges (e.g.,within 10-20% deviation from a base frequency such as a range of 9 GHzto 11 GHz) when used in isolation. Ring-based PLL have significantlywider tuning ranges (e.g., 800 MHz to 6 GHz) but often suffer from poorphase noise when used in isolation. In the cascaded configurationdescribed above, the wide range of tuning frequencies associated withring-based PLL 120 can be achieved at output signal 106. Furthermore,input 104 to ring-based PLL 120 can be configured to have a bandwidth(e.g., 1 GHz or higher) that is high enough that most of ring PLL's 120phase noise is filtered out. Furthermore, periodic phase noise of theLC-based PLL 110 due to fractional divider and charge pump mismatch canbe canceled or reduced via feedforward control circuit 130 as will bedescribed in further detail below with respect to FIGS. 4-5. As aresult, cascaded PLL architecture 100 can achieve both good phase noiseand a wide tuning range (e.g., 800 MHz to 6 GHz) for compliance with alarge variety of wireless standards.

In an alternative embodiment, a cascaded PLL architecture includes anLC-based PLL 110 coupled to a ring-based PLL 120, but omits thefeedforward control path (feedforward control 130 and signals 112, 114)shown in FIG. 1. Omitting the feedforward control path may result inincreased phase noise, but the architecture can still beneficiallyachieve a wide tuning range and good enough noise performance that itmay be suitable for some applications. In another embodiment, differenttypes of PLLs may be used as first PLL 110 and second PLL 120 (with orwithout the feedforward control path). For example, in variousembodiments, first PLL 110 and second PLL 120 can each either comprise afractional-N PLL or an integer-N PLL. Furthermore, first PLL 110 andsecond PLL 120 can each either comprise an LC-based PLL or a ring-basedPLL. In other alternative embodiments, three or more cascaded PLLs maybe used with at least one feedforward path between them for noisecancellation.

LC-based Phase Locked Loop Circuit

FIG. 2 is a circuit diagram illustrating an LC-based PLL circuit 200that could be used in the cascaded PLL configuration described above orcould be used in a standalone configuration, according to an embodiment.Phase frequency detector (PFD) 202 receives input signal 204 having areference frequency f_(REF) and PLL feedback signal 206 having afeedback frequency f_(FB). PFD 202 detects the difference in phase andfrequency between input signal 204 and feedback signal 206 and generatesphase difference signal 208 indicating whether feedback signal 206 lagsor leads input signal 204. For example, in one embodiment, phasedifference signal 208 comprises an “up signal” (e.g., a logic highsignal) if feedback signal 206 leads input signal 204 (indicating thatthe PLL frequency should be increased) and phase difference signal 208comprises a “down signal” (e.g., a logic low signal) if feedback signal206 lags input signal 204 (indicating that the PLL frequency should bedecreased).

Control element 210 receives phase difference signal 208 and generatesfrequency control signal 224 for controlling variable frequencyoscillator 226. In one embodiment, control element 210 comprises chargepump 212 and loop filter 222. Charge pump 212 drives current into ordraws current from loop filter 222 based on phase difference signal 208.In one embodiment, charge pump 212 is implemented as a first currentsource 214 coupled to a first switch 216 and a second current source 218coupled to a second switch 220. When phase difference signal 208indicates an “up signal,” first switch 216 turns on, thereby couplingfirst current source 214 to loop filter 222, and second switch 220 turnsoff, thereby decoupling second current source 218 from loop filter 222.This configuration causes a positive current to flow through loop filter222 and the voltage of frequency control signal 224 increases.Alternatively, when phase difference signal 208 indicates a “downsignal,” first switch 216 turns off, thereby decoupling first currentsource 214 from loop filter 222, and second switch 220 turns on, therebycoupling second current source 218 to loop filter 222. Thisconfiguration causes a negative current to flow through loop filter 222and the voltage of frequency control signal 224 decreases. Loop filter222 filters out jitter and reduces voltage overshoot when charge pump212 switches between the up configuration and the down configuration.For example, in one embodiment, the loop filter is implemented as apassive RC filter. In alternative embodiments, a different configurationof control element 210 may be used to generate frequency control signal224 from phase difference signal 208.

Variable frequency oscillator 226 receives the frequency control signaland generates an oscillating output signal 228 having an outputfrequency f_(PLL), that varies based on frequency control signal 224.Variable frequency oscillator 226 may be implemented as, for example, anLC voltage controlled oscillator (VCO). In general, the frequencyf_(PLL), of output signal 228 will have a fractional-N relation with thereference frequency f_(REF), as will be explained in further detailbelow.

Circuit element 230 generates first phase signal 232 and second phasesignal 234 based on output signal 228. First phase signal 232 and secondphase signal 234 have the same frequency but second phase signal 234 isphase-shifted relative to first phase signal 232. In one embodiment,circuit element 230 comprises a frequency divider (e.g., a dividefrequency by two circuit) and first phase signal 232 and second phasesignal 234 comprise in-phase (I) and quadrature phase (Q) componentsignals respectively each having a frequency of f_(PLL)/2.Alternatively, a different phase-shift may be applied and first phasesignal 232 and second phase signal 234 are not necessarily 90 degreesout of phase.

Phase interpolator 236 receives first phase signal 232 and second phasesignal 234 and generates interpolated signal 244 based on a modulatedphase control signal 238. Interpolated signal 244 comprises a signalhaving an average phase in between (or equal to) the phases of firstphase signal 232 and second phase signal 234.

Frequency divider circuit 246 receives interpolated signal 244 anddivides the frequency of interpolated signal 244 to generate PLLfeedback signal 206. For example in one embodiment, frequency dividercircuit 246 divides by N/2, where N is predefined integer value.

In one embodiment, frequency selection signal 242 provides a desiredfrequency to control output frequency 228 of PLL 200. Modulator 240(e.g., a delta-sigma modulator) receives frequency selection signal 242that generates modulated phase control signal 238. In one embodiment,modulated phase control signal 238 is based on a high frequency clockhaving a frequency much higher than the frequency of first phase signal232 and second phase signal 234. Modulated phase control signal 238specifies a selected phase for each period of modulated phase controlsignal 238 from two or more selectable phases. Modulator 240 selectsbetween the possible phases such that an average phase over time isadded by phase interpolator 236 to achieve the desired frequencyspecified by frequency selection signal 242. For example, in oneembodiment, modulator 240 applies a delta-sigma modulation technique toachieve the correct average phase.

In one embodiment, the average phase of modulated signal 244 (relativeto the first phase signal 232) is controllable based on a parameter Pand a parameter L of phase interpolator 236, where 2^(P)−1 specifies aprogrammable step size out of 2^(L) phase steps evenly spaced between 0°and 360°. For example, if L=4, P=2, there are 16 phase steps (e.g., 0°,22.5° 45°,67.5°, 90°, . . . ) and phase interpolator 236 adds, onaverage, a phase of 3 step sizes (67.5°) to first phase signal 232.Applying a phase shift at each cycle is equivalent to a fractionalincrease in period, thereby achieving a fractional decrease in frequencyof modulated signal 244 relative to first phase signal 232.

The feedback loop of PLL circuit 200 operates to configure the outputfrequency f_(PLL) so that once divided/reduced in frequency, thefeedback frequency f_(FB) will match the reference frequency f_(REF). Byvarying the parameters, P, L, and N, a variety of different frequenciescan be achieved at the output f_(PLL), from a single reference frequencyf_(REF).

Operation of PLL circuit 200 can be further understood in view of theequations below. As stated above, the desired output frequency f_(PLL)is achieved when the feedback frequency f_(FB) matches the referencefrequency f_(REF), or equivalently, the reference period T_(REF) matchesthe feedback period T_(FB):

T _(REF) =T _(FB)  (1)

The feedback period T_(FB) is given by:

$\begin{matrix}{T_{FB} = {\lbrack {{2T_{PLL}} \pm {\frac{2T_{PLL}}{2^{L}}( {2^{P} - 1} )}} \rbrack \frac{N}{2}}} & (2)\end{matrix}$

where T_(PLL) is the period of output signal 228 (therefore 2T_(PLL) isthe period of first phase signal 232 and second phase signal 234generated by element 230), 2 ^(L) is the number of average phase steps(evenly spaced between 0° and 360°) that phase interpolator 236 can addto the first phase signal 232, 2^(P)−1 is the average number of stepsapplied, and N/2 is the frequency division ratio of frequency divider246. Thus, as can be seen from equation (2), phase interpolator 236operates to lengthen or reduce the period 2T_(PLL) of first phase signal234 by some fractional amount. The period is then further lengthened byN/2 to enable a wide range of periods having fraction relationships tothe reference period T_(REF). Solving equations (1) and (2) for T_(REF)yields:

$\begin{matrix}{T_{REF} = {\lbrack {1 \pm \frac{2^{P} - 1}{2^{L}}} \rbrack {NT}_{PLL}}} & (3)\end{matrix}$

Converting equation (3) to frequency gives:

$\begin{matrix}{f_{PLL} = {{N\lbrack {1 \pm \frac{2^{P} - 1}{2^{L}}} \rbrack}f_{REF}}} & (4)\end{matrix}$

where f_(PLL)=1/T_(PLL) and f_(REF)=1/T_(REF). As can be seen fromequation (4), a desired output frequency f_(PLL), can be achieved from areference frequency f_(REF) by varying parameters N, P, and L.

FIG. 3 illustrates example waveforms showing operation of an exampleLC-based PLL circuit. Waveform 302 illustrates an example of a modulatedsignal (e.g., signal 244) produced by a phase interpolator controlled bya modulator. As can be seen, the phase interpolator produces a modulatedoutput signal having an average phase of Φ_(avg) relative to a referencephase. To achieve this average phase, output of the phase interpolatoris time modulated between three possible phases Φ₁, Φ₂, and Φ₃ inaccordance with a delta-sigma modulation technique. Waveform 304illustrates an example of a frequency control signal (e.g., signal 224)applied to an LC-based voltage controlled oscillator. As can be seen,the voltage V_(Control) is stepped up or down over time to providedithering around an average voltage that will achieve the desired outputfrequency of the PLL. This dithering results in phase jitter at theoutput of the PLL as can be seen in waveform 306. The phase jitterrepresents a deviation in time from ideal zero-crossings of the outputsignal (e.g., in relation to an ideal clock of the desired outputfrequency) and results from quantization error of the PLL circuit. Thisjitter results in typically undesired spur noise in the frequencyspectrum of the output signal of the PLL circuit. However, such spurnoise can be reduced or canceled in the cascaded PLL configuration aswill be explained in further detail below.

FIG. 4 is a more detailed circuit diagram of an embodiment of a cascadedPLL circuit 400 illustrating how spur noise cancellation can beachieved. In this embodiment, first PLL circuit 410 comprises an LC VCOsimilar or identical in architecture and functionality as PLL circuit200 of FIG. 2. For example, in one embodiment, first PLL circuit 410produces an output having a frequency given by

$f_{INT} = {{N\lbrack {1 \pm \frac{2^{P} - 1}{2^{L}}} \rbrack}{f_{REF}.}}$

In second PLL circuit 420, frequency divider 444 receives intermediatesignal 442 having a frequency f_(INT) outputted by first PLL circuit410. In one embodiment, frequency divider 444 divides frequency ofintermediate signal 442 by an integer M to produce signal 446 having afrequency f_(INT)/M. Phase-frequency detector 448 detects a differencein phase and frequency between signal 446 and feedback signal 450 andproduces phase difference signal 452 (which may be, for example, an “upsignal” or a “down signal” as described above). Control element 454operates similarly to control element 210 described above except thatsecondary current sources 456, 458 are included which are controlled byfeedforward signal 414 from feedforward control circuit 430. Note thatalthough secondary current sources 456, 458 are illustrated ascurrent-controlled current sources, these could alternatively beimplemented as voltage-controlled current sources by omitting block 434.Thus, some additional current is driven into or drawn from the loopfilter 472 based on feedforward control signal 414, operation of whichwill be described in further detail below. Note that in practice, secondcurrent sources 456, 458 could be, but are not necessarily separatephysical devices from fixed current sources 457, 459.

Control element 454 produces frequency control signal 460 that controlsoutput of ring voltage-controlled oscillator (VCO) 462. The VCO producesoutput signal 464 having a frequency f_(OUT) which is based on frequencycontrol signal 462. In a feedback loop, frequency divider 466 receivesoutput signal 464 and divides its frequency by an integer A to generatefeedback signal 450. Thus, second PLL circuit 420 operates as aninteger-N PLL with an output frequency

$f_{OUT} = {\frac{A}{M}{f_{INT}.}}$

The overall transfer function of cascaded PLL circuit 400 is thereforegiven by

$f_{OUT} = {\frac{A}{M}{N\lbrack {1 \pm \frac{2^{P} - 1}{2^{L}}} \rbrack}{f_{REF}.}}$

Therefore, the overall cascaded PLL circuit 400 operates as afractional-N PLL because the output frequency f_(OUT) can have afractional relationship with the reference frequency f_(REF).

Feedforward control circuit 430 receives control signal 412 from firstPLL circuit 410 and generates feedforward signal 414. In one embodiment,control signal 412 comprises a frequency control signal used to controlan LC VCO of first PLL circuit 410. As can be seen in the waveforms ofFIG. 3, this signal 412 is representative of the phase jitter that willappear at intermediate output signal 442. In one embodiment, feedforwardcontrol circuit 430 comprises a transfer function block 432 and avoltage-to-current block 434. Transfer function block 432 applies atransfer function H(s) to control signal 412 where s is frequency in theLaplace domain. Voltage-to-current block 434 converts the voltagecontrol signal from the transfer function block 432 to generate acurrent control signal (feedforward signal 414) for controlling currentof secondary current sources 456, 458 of control element 454 of secondPLL circuit 420. Transfer function block 432 is configured to apply atransfer function H(s) describing the negative relationship betweencontrol signal 412 and the resulting jitter appearing at intermediateoutput signal 442, so that the jitter appearing at output 442 of firstPLL circuit 410 is canceled or minimized in second PLL circuit 420.

FIG. 5 illustrates a phase-domain linear representation of a cascade PLLcircuit (such as the cascaded PLL circuit 400 of FIG. 4) showingoperation of a feedforward control path for spur and noise cancellation.Input signal 504 has an input phase Φ_(IN). Phase adder block 502computes a difference in phase between the input phase Φ_(IN) and afeedback phase Φ_(FB) of a feedback signal 506. Gain block 552 applies again K_(PD1) in the phase domain and produces output 508 representing anamplified difference in the phase domain. Blocks 502, 552 collectivelyapply a transfer function in the phase domain equivalent to that ofphase-frequency detector 202 of FIG. 2. Blocks 510 and block 511 applyphase gains of K_(P1) and K₁₁/s respectively to signal 508 and theoutputs from blocks 510, 511 are added by phase adder 516, where srepresents frequency in the Laplace domain. Blocks 510, 511, and 516collectively function as a phase integrator with some gain and modelcontrol element 210 in the phase domain to produce control voltage 512.Block 526 applies a phase gain K_(VCO1)/s to produce the intermediateoutput signal which has an intermediate output phase Φ_(INT). Block 526models variable frequency oscillator 226 of FIG. 2. In block 532, aphase gain of ½ is applied which models frequency divider 230 in thephase domain. Phase adder 536 adds a phase Φ_(PI) and models phaseinterpolator 236. Block 546 applies a phase gain of 2/N which representsfrequency divider 246. Phase adder 548 represents phase noise Φ_(N) thatis added based on quantization error of the phase interpolator. Thisphase noise Φ_(N) will propagate through the circuit and will result inspur noise unless canceled.

In the linear representation of the second PLL, phase gain block 544applies a phase gain of 1/M, modeling block 444 of FIG. 4. Phase adderblock 548 and phase gain block 592 collectively represent thephase-frequency detector 448. Phase gain blocks 558, 560 and phase adderblocks 556, 564 collectively model element 454. Particularly, phaseadder 556 reduces phase based on the feedforward current 514 produced bysecondary current sources 456, 458. Gain block 592 applies someadditional gain K_(PD2) in the phase domain, to produce an output signalhaving an output phase Φ_(OUT) and represents ring VCO 462. Phase gainblock 566 represents frequency divider 466.

To cancel propagation of the phase noise Φ_(N) to the output phaseΦ_(OUT), a transfer function H_(QC) is applied to control voltage 512 togenerate the feedforward current 514. Absent the cancellation pathincluding block 530, quantization noise Φ_(N) would be transferred tothe output phase Φ_(OUT) according to the transfer function:

$\begin{matrix}{{H_{PLL}(s)} = \frac{{H_{{PLL}\; 1}(s)}{H_{{PLL}\; 2}(s)}}{M}} & (5)\end{matrix}$

where H_(PLL1)(s) is the transfer function of the first PLL from inputsignal 504 having phase Φ_(IN) to intermediate output signal 542 havingphase Φ_(INT); and H_(PLL2)(s) is the transfer function of the secondPLL (excluding frequency divider block 544) from signal 572 to outputsignal 564 having a phase Φ_(OUT). Furthermore, the phase noise Φ_(N)will propagate through the cancellation path (represented by block 530)according to the transfer function:

$\begin{matrix}{{H_{C}(s)} = {{- {H_{QC}(s)}}\frac{H_{{PLL}\; 2}(s)}{K_{{PD}\; 2}}\frac{{sH}_{{PLL}\; 1}(s)}{K_{{VCO}\; 1}}}} & (6)\end{matrix}$

Thus, for noise cancellation, the following condition should be met:

$\begin{matrix}{\frac{{H_{{PLL}\; 1}(s)}{H_{{PLL}\; 2}(s)}}{M} = {{H_{QC}(s)}\frac{H_{{PLL}\; 2}(s)}{K_{{PD}\; 2}}\frac{{sH}_{{PLL}\; 1}(s)}{K_{{VCO}\; 1}}}} & (7)\end{matrix}$

Solving for H_(QC) provides an appropriate transfer function for spurcancellation:

$\begin{matrix}{H_{QC} = {\frac{K_{{PD}\; 2}K_{VCO}}{M}\; 1\text{/}s}} & (8)\end{matrix}$

Observing that a capacitor with a capacitance C has a Laplace transformof 1/sC, a capacitor can be used to implement block 530 (or block 430 inFIG. 4). The value of the capacitor for spur cancellation is given by:

$\begin{matrix}{C = \frac{M}{K_{{PD}\; 2}K_{VCO}}} & (8)\end{matrix}$

PLL with Split-Tuned Ring VCO

FIG. 6A illustrates an embodiment of a ring-based PLL 600 with a splittuned ring VCO. PLL 600 is similar in architecture and function to PLLcircuit 420 of FIG. 4 but generates two output signals 664, 668 (e.g., Iand Q signals) and includes a split-tuned VCO 662 with a phasecorrection loop. The PLL circuit 600 could be used in a standaloneconfiguration or cascaded with another PLL circuit as described above.For example, the PLL circuit 600 could be used as the second PLL 120 ofFIG. 1.

The two output signals 664, 668 comprise signals having the samefrequency f_(OUT) but different phases. In one embodiment, for example,the first output signal 664 is an in-phase signal I and the secondoutput signal 668 comprises a quadrature signal Q having a 90° phaseshift relative to the first output signal 664. Alternatively, the secondoutput signal 668 could have a different fixed phase relationship to thefirst output signal 664 that is not necessarily 90°.

PLL circuit 600 comprises a phase-frequency detector 648 that generatesphase difference signal 652 based on a difference in phase between inputsignal 646 and feedback signal 650. Control element 654 generatesfrequency control signal 660 based on phase difference signal 652. Inone embodiment, for example, control element 654 comprises a charge pumpand a loop filter as described above. Variable frequency oscillator 662generates first output signal 664 and second output signal 668 based onfrequency control signal 660 and oscillator feedback signal 670.Furthermore, variable frequency oscillator 662 controls the relativephases of first output signal 664 and second output signal 668 based onoscillator feedback signal 670 as will be described in further detailbelow. Error detection module 676 detects a difference in phase betweenfirst output signal 664 and second output signal 668 and compares thedetected phase difference to a desired phase difference to determine aphase error. Error detection module 676 then generates oscillatorfeedback signal 670 indicative of such phase error to reduce thedetected phase error. Frequency divider circuit 666 divides thefrequency of second output signal 668 (or alternatively, first outputsignal 664) to generate feedback signal 650.

Ring VCO 662 comprises a chain of circuit elements each with some finitepropagation delay such as, for example, a chain of inverters and/ornon-inverting buffers. For example, in one embodiment, ring VCO 662comprises an odd number of inverters. A finite amount of time after aparticular logic level is applied to the first input, the last inverterin the chain outputs the inverse logic level. This output is fed back tothe input, thus causing an oscillation with a frequency based on theoverall delay through ring oscillator 662. The circuit elements of ringoscillator 662 have a controllable delay, thus enabling variousoscillation frequencies to be achieved.

In the illustrated embodiment, a first set of circuit elements (e.g.,inverters and/or non-inverting buffers) of ring oscillator 662 aregrouped together as first delay element 672 with their delays controlledby control signal 660. A second group of circuit elements (e.g.,inverters and/or non-inverting buffers) are grouped together as seconddelay element 674 with their delays controlled by oscillator feedbacksignal 670. In one embodiment, first delay element 672 and second delayelement 674 have different numbers of inverters to ensure an odd numberof overall inverters (e.g., first delay element 672 has an odd number ofinverters and second delay element 674 has an even number of inverters)in ring oscillator 662. Thus, the overall frequency of VCO 662 isdetermined by the combined delay through first and second delay elements672, 674. The phase difference between first output signal 664 andsecond output signal 668 is determined based on the difference in delaysbetween first delay element 672 and second delay element 674. Thus, abenefit of using a ring oscillator 662 is that multiple output signals664, 668 having different phases can be drawn from oscillator 662without requiring an additional phase shifting element separate from theoscillator.

Error detection module 676 detects the phase delay of second outputsignal 668 relative to first output signal 664. Error detection module676 then compares the phase delay to a desired phase delay. If thedetected phase delay is greater than desired (i.e., second output signal668 lags too far behind first output signal 664), error detection module676 adjusts oscillator feedback signal 670 to decrease the delay throughsecond delay element 674. This will momentarily cause an overallincrease in frequency of first and second output signals 664, 668.However, the PLL circuit 600 will compensate for the frequency increaseby adjusting control signal 660 to cause a corresponding increase indelay of first delay element 672, thereby maintaining the desired outputfrequency. Similarly, when the detected phase delay is less than desired(i.e., second output signal 668 does not lag far enough behind firstoutput signal 664), error detection module 676 will cause an increase indelay through second delay element 674, and the PLL circuit 600 willcause a corresponding decrease in delay through first delay element 672to achieve both the desired output frequency and the desired phasedifference between first and second output signal 664, 668.

FIG. 6B illustrates an embodiment of a split-tuned LC-based VCO 680 thatcan be used in the place of the split-tuned ring VCO 662 show in FIG.6A. The split-tuned LC-based VCO 680 receives frequency control signal660 and oscillator feedback signal 670, and produce first and secondoutput signals 664, 668. In one embodiment, the split-tuned LC-based VCOcomprises a first LC-based VCO 682, a second LC-based VCO 686, a firstcoupling element 684, and a second coupling element 688. The firstLC-based VCO 686 receives a first reference signal 694 having a firstreference frequency and generates a first output signal 668 having afirst output frequency based on frequency control signal 660. Firstcoupling element 684 receives the first output signal 668 having a phaseφ and applies a phase shift by subtracting a phase Δ to generate signal692, where the phase shift amount Δ is based on oscillator feedbacksignal 670. Second LC-based VCO 686 receives signal 692 (a secondreference signal) and generates a second output signal 664 having asecond output frequency based on frequency control signal 660. Secondcoupling element 688 receives second output signal 664 having a phase φand applies a phase shift by adding the phase Δ to generate signal 694,where the phase shift amount Δ is based on oscillator feedback signal670 and is the same as shift amount applied by coupling element 684.Signal 694 outputted by second coupling element 688 is the input tofirst LC-based VCO 682, thus forming a closed loop system. In oneembodiment, first and second output signals 668, 664 are Q and I signalsrespectively of a quadrature output and are thus 90° out of phase.

An example circuit for use as the first LC-based VCO 682 is illustratedcomprising inductors L1, L2, transistors M1, M2, M3, M4, current source696, and variable capacitors C1, C2 arranged as an LC-based VCO.Frequency control input 660 controls capacitance of variable capacitorsC1, C2 in order to achieve variations in frequency of the output signal668 (which is shown as a differential signal) in relation to thereference signal 694 (which is shown as a differential signal). SecondLC-based VCO 686 may have a similar or identical architecture. Inalternative embodiments, different variations of LC-based VCOs can beused that operate according to similar principles.

An example circuit for coupling element 684 is also illustratedcomprising resistors R1, R2 and transistors M5-M10. The circuit elementsare arranged to operate as a phase shifter to shift a phase of an input(e.g., first output signal 668 which is shown as a differential signal)based on an amount Δ proportional to the difference between V+ and V−,to produce an output (e.g., signal 692 which is shown as a differentialsignal). Coupling element 688 can be implemented according to similar orthe same architecture as coupling element 684 except that differentialinputs V+ and V− are switched in order to achieve a shift of +Δ insteadof −Δ. In alternative embodiments, different variations of couplingelements 684, 688 can be used that operate according to similarprinciples.

The LC-based VCO 680 achieves a similar function to the ring-based VCO662 described above. Frequency control signal 660 controls overalloscillation frequency of the LC-based VCO 680. Feedback oscillationsignal 670 controls an amount of phase shift between the two outputsignals 668, 664 in order to achieve the desired phase relationship(e.g., quadrature signals that are 90° out of phase).

FIG. 7 illustrates one example implementation of an error detectionmodule 766 for use in a phase correction loop for a split-tuned VCO 762which could be, for example, a split-tuned ring-based VCO 662 asdescribed in FIG. 6A or a split-tuned LC-based VCO 680 as described inFIG. 6B. The error detection module 766 could be used as error detectionmodule 676 in FIG. 6A described above. Frequency divider 772 and phaseinterpolator 774 generate phase rotating clock signal 776 based on inputclock signal 760 having a same frequency f_(OUT) as output signals 764,768. Specifically, frequency divider 772 divides input clock signal 760into two signals 778, 780 having different phases (e.g., two signalsthat are 90° out of phase). Phase interpolator 774 then interpolates thetwo signals 778, 780, where the mixture of signals 778, 780 changes overtime. For example, in one embodiment, the amount of phase shift appliedto signal 778 increases incrementally over time to achieve a rotatingphase of phase rotating clock signal 776. Sampler 782 samples firstoutput signal 764 and second output signal 768 based on rotating phaseclock signal 776 and the samples are provided to digital processingmodule 784. By determining transition timing of the samples (e.g., whenthe samples transition from low to high or high to low), digitalprocessing module 784 can determine the relative timing of thetransitions and therefore determine the phase difference between firstinput signal 764 and second input signal 768. Digital processing module784 outputs phase error signal 786 representing the detected differencein phase. Gain blocks 790, 788 and adder 792 collectively implement adigital filter for filtering phase error signal 786. Alternatively, ananalog filter or a different type of digital filter implementation couldbe used. The filtered signal is outputted to modulator 796 (e.g., adelta-sigma modulator) and digital-analog converter 798 to generateoscillator feedback signal 770. Modulation provides quantization noiseshaping in the phase correction loop to reduce noise based on limitedresolution of the digital phase detection.

In other alternative embodiments, different combinations of theabove-described PLL circuits may be used in standalone configurations orin cascaded configurations. Furthermore, one or more of the abovedescribed PLL circuits can be used in a cascaded configuration with oneor more conventional PLL architectures. Beneficially, the describedembodiments enable a wide range of frequencies to be synthesized from asingle oscillator with high noise performance, thereby enablingcompliance with a wide variety of wireless communication standards.

Processes Performed by the PLL Circuits

FIGS. 8A-D illustrate examples of processes that can be performed by thevarious embodiments of the PLL circuits described above. FIG. 8Aillustrates an embodiment of a process performed by a cascaded PLLcircuit having an architecture such as that described with respect toFIG. 1 above. In this process, a first phase-locked loop circuitreceives 802 a reference signal having a first frequency. The firstphase phase-locked loop circuit generates 804 a control signal for afeedforward path and an intermediate signal having a second frequency. Afeedforward circuit generates 806 a feedforward signal based on thecontrol signal from the first phase-locked loop. A second phase-lockedloop circuit cascaded with the first phase-locked loop circuit generates808 an output signal signal having a third frequency based on theintermediate signal and the feedforward signal. This feedforward pathcan beneficially reduce or eliminate propagation of phase noise from thefirst phase-locked loop circuit to the output signal.

FIG. 8B illustrates another process that can be performed by a cascadedPLL circuit having an architecture such as that described above withrespect to FIG. 1. In this process, a first phase-locked loop circuitreceives 812 a reference signal having a first frequency. The firstphase-locked loop circuit generates 814 an intermediate signal having asecond frequency using an LC-based variable frequency oscillator. Asecond phase-locked loop circuit generates 816 an output signal having athird frequency based on the intermediate signal using a ring-basedoscillator.

FIG. 8C illustrates a process that can be performed by a PLL circuithaving an architecture such as that described above with respect to FIG.2. In this process, a phase detector circuit receives 822 an inputsignal and generates 824 a phase difference signal based on a differencein phase between the input signal and a phase-locked loop feedbacksignal. A control element generates 826 a frequency control signal basedon the phase difference signal. A variable frequency oscillatorgenerates 828 a first output signal and a second output signal in whichthe second output signal is phase-shifted relative to the first outputsignal. Here, the frequency of the first and second output signals iscontrolled based on the frequency control signal and an oscillatorfeedback signal, and a phase of the second output signal relative to thefirst output signal is controlled based on the oscillator feedbacksignal. A phase correction circuit determines 830 a phase error betweena detected difference in phases of the first and second output signalsand a desired difference in phases of the first and second outputsignals and generates 832 the oscillator feedback signal to reduce thedetected phase error. A frequency divider circuit divides 834 afrequency of the first output signal or the second output signal togenerate the phase-locked loop feedback signal.

FIG. 8D illustrates another process that can be performed by a PLLcircuit having an architecture such as that described above with respectto FIG. 2. In this process, a phase detector receives 842 an inputsignal having a reference frequency and generates 844 a phase differencesignal based on a difference in phase between an input signal having areference frequency and a phase-locked loop feedback signal. A controlcircuit generates 846 a frequency control signal based on the phasedifference signal. A variable frequency oscillator generates 848 anoutput signal having a frequency adjusted based on the frequency controlsignal and having a fractional-N relation with the reference frequency.A circuit element generates 850 a first phase signal and a second phasesignal based on the output signal. Here, both the first and second phasesignals have the same frequency as the output signal, and the secondphase signal is phase-shifted relative to the first phase signal. Amodulator generates 852 a modulated phase control signal specifying aselected phase for each period of the modulated phase control based on afrequency selection signal. A phase interpolator generates 854 aninterpolated signal for each period of the modulated phase controlsignal, the interpolated signal having the selected phase selected froma plurality of selectable phases in between phases of the first phasesignal and the second phase signal. A frequency divider circuit thendivides 856 a frequency of the interpolated signal to generate thephase-locked loop feedback signal.

In other alternative embodiments, different variations of the exampleprocesses described above can be performed by the described frequencysynthesizers in order to synthesize signals. For example, in variousembodiments, the process steps of FIGS. 8A-8D can be performed in ordersother than the orders shown or steps can be performed in parallel.Furthermore, the process steps can be performed by components other thanthose described.

Upon reading this disclosure, those of ordinary skill in the art willappreciate still alternative structural and functional designs for afrequency synthesizer and processes for frequency synthesis, through thedisclosed principles of the present disclosure. Thus, while particularembodiments and applications of the present disclosure have beenillustrated and described, it is to be understood that the disclosure isnot limited to the precise construction and components disclosed herein.Various modifications, changes and variations which will be apparent tothose skilled in the art may be made in the arrangement, operation anddetails of the method and apparatus of the present disclosure hereinwithout departing from the scope of the disclosure as defined in theappended claims.

What is claimed is:
 1. A circuit comprising: a first phase-locked loopcircuit to receive a reference signal having a first frequency andgenerate an intermediate signal having a second frequency; a feedforwardcircuit to receive a control signal from the first phase-locked loopcircuit and generate a feedforward signal; and a second phase-lockedloop circuit to receive the intermediate signal having the secondfrequency and the feedforward signal, and generate an output signalhaving a third frequency based on the intermediate signal and thefeedforward signal.
 2. The circuit of claim 1, wherein the feedforwardsignal represents phase noise in the intermediate signal having thesecond frequency, and wherein the second phase-locked loop circuitreduces propagation of the phase noise from the intermediate signal tophase noise of the output signal based on the feedforward signal.
 3. Thecircuit of claim 1, wherein the first phase-locked loop circuitcomprises a fractional-N phase-locked loop circuit and wherein thesecond phase-locked loop circuit comprises an integer-N phase lockedloop circuit.
 4. The circuit of claim 1, wherein the first phase-lockedloop circuit comprises an LC voltage controlled oscillator and whereinthe second phase-locked loop circuit comprises a ring voltage controlledoscillator.
 5. The circuit of claim 1, wherein the feedforward circuithas a transfer function based on a phase gain of a voltage controlledoscillator of the first phase-locked loop circuit, a phase gain of aphase detector of the second phase-locked loop circuit, and a forwardfrequency divider ratio of the second phase-locked loop circuit.
 6. Thecircuit of claim 1, wherein the feedforward circuit has a transferfunction based on a negative relationship between a voltage applied to avoltage controlled oscillator of the first phase-locked loop circuit andjitter of the intermediate output signal.
 7. The circuit of claim 1wherein the first phase-locked loop comprises: a phase detector togenerate a phase difference signal based on a difference in phasebetween the input signal having the first frequency and a phase-lockedloop feedback signal; a control element to generate the control signalbased on the phase difference signal; a variable frequency oscillator togenerate the intermediate signal having a frequency adjusted based onthe control signal; a circuit element to generate a first phase signaland a second phase signal based on the intermediate signal, both thefirst and second phase signals having a same frequency as theintermediate signal, and the second phase signal phase-shifted relativeto the first phase signal; a modulator to receive a frequency selectionsignal and generate a modulated phase control signal specifying aselected phase for each period of the modulated phase control signal; aphase interpolator to receive the modulated phase control signal and thefirst and second phase signals, the phase interpolator outputting aninterpolated signal for each period of the modulated phase controlsignal, the interpolated signal having the selected phase selected froma plurality of selectable phases in between phases of the first phasesignal and the second phase signal; and a frequency divider circuit todivide a frequency of the interpolated signal and generate thephase-locked loop feedback signal.
 8. The circuit of claim 1, whereinthe second phase-locked loop circuit comprises: a forward frequencydivider circuit to divide a frequency of the intermediate signal fromthe first phase-locked loop circuit and generate a frequency-dividedinput signal; a phase detector to generate a phase difference signalbased on a difference in phase between the frequency-divided inputsignal and a phase-locked loop feedback signal; a control element togenerate a frequency control signal based on the phase difference signaland the feedforward signal; a variable frequency oscillator to generatethe output signal based on the frequency control signal; and a feedbackfrequency divider circuit to divide a frequency of the output signal andgenerate the phase-locked loop feedback signal.
 9. The circuit of claim8, wherein the control element comprises: a charge pump to provide acurrent based on the phase difference signal and the feedfoward signal;and a loop filter to produce the frequency control signal based on thecurrent.
 10. The circuit of claim 8, wherein the variable frequencyoscillator furthermore generates a quadrature signal having a phaseshift relative to the output signal based on an oscillator feedbacksignal, the second phase-locked loop circuit further comprising: anerror correction loop to determine a phase error between a detecteddifference in phases of the output signal and the quadrature signal, tocompare the detected difference to a desired difference in phases, andto generate the oscillator feedback signal to reduce the detected phaseerror.
 11. A circuit comprising: a phase detector to generate a phasedifference signal based on a difference in phase between an input signaland a phase-locked loop feedback signal; a control element to generate afrequency control signal based on the phase difference signal; avariable frequency oscillator to generate a first output signal and asecond output signal, the second output signal phase-shifted relative tothe first output signal, the variable frequency oscillator to control afrequency of the first and second output signals based on the frequencycontrol signal and an oscillator feedback signal, and the variablefrequency oscillator controlling a phase of the second output signalrelative to the first output signal based on the oscillator feedbacksignal; and a phase correction circuit to determine a phase errorbetween a detected difference in phases of the first and second outputsignals and a desired difference in phases of the first and secondoutput signals, and generate the oscillator feedback signal to reducethe detected phase error; and a frequency divider circuit to divide afrequency of the first output signal or the second output signal andgenerate the phase-locked loop feedback signal.
 12. The circuit of claim11, wherein the variable frequency oscillator comprises a ring-basedvariable frequency oscillator including: a first variable delay elementto apply a first delay to the first output signal based on the frequencycontrol signal and generate the second output signal; and a secondvariable delay element coupled in series with the first variable delayelement, the second variable delay element to apply a second delay tothe second output signal based on the oscillator feedback signal andgenerate the first output signal.
 13. The circuit of claim 12, whereinthe phase correction circuit controls the second variable delay elementso that the second delay is substantially equal to the first delay. 14.The circuit of claim 11, wherein the variable frequency oscillatorcomprises an LC-based variable frequency oscillator including: a firstLC-based voltage controlled oscillator to generate the first outputsignal based on a first reference signal and the frequency controlsignal; a first coupling element to receive the first output signal andapply a first phase shift based on the oscillator feedback signal togenerate a second reference signal; a second LC-based voltage controlledoscillator to generate the second output signal based on the secondreference signal and the frequency control signal; and a second couplingelement to receive the second output signal and apply a second phaseshift based on the oscillator feedback signal to generate the firstreference signal.
 15. The circuit of claim 14, wherein the first phaseshift comprises subtracting a given phase amount from the first outputsignal based on the oscillator feedback signal and wherein the secondphase shift comprises adding the phase amount to the second outputsignal based on the oscillator feedback signal.
 16. The circuit of claim11, wherein the phase correction circuit comprises: a time-to-digitalconverter to sample the first output signal and the second output signaland generate a digital code representing a phase difference between thefirst and second output signals, wherein the oscillator feedback signalis generated based on the digital code.
 17. The circuit of claim 16,wherein the time-to-digital converter comprises: a phase interpolator togenerate a phase-rotating clock signal; and a sampler for sampling thefirst and second output signals based on the phase-rotating clocksignal.
 18. The circuit of claim 16, wherein the phase correctioncircuit further comprises: a modulator to modulate the digital code fromthe time-to-digital converter based on a high frequency clock togenerate the oscillator feedback signal, the high frequency clock havinga frequency higher than the frequency of the first and second outputsignal, wherein the oscillator feedback signal is modulated to achievean average output representing the digital code.
 19. A method forsynthesizing an output signal, the method comprising: receiving, by afirst phase-locked loop circuit, a reference signal having a firstfrequency; generating, by the first phase-locked loop circuit, a controlsignal and an intermediate signal having a second frequency; generating,by a feedforward circuit, a feedforward signal based on the controlsignal from the first phase-locked loop circuit; generating, by a secondphase-locked loop circuit, the output signal having a third frequencybased on the intermediate signal having the second frequency and thefeedforward signal; and outputting the output signal having the thirdfrequency.
 20. The method of claim 19, wherein generating the outputsignal comprises: applying a transfer function to the intermediatesignal based on a phase gain of a voltage controlled oscillator of thefirst phase-locked loop circuit, a phase gain of a phase detector of thesecond phase-locked loop circuit, and a forward frequency divider ratioof the second phase-locked loop circuit.
 21. The method of claim 19,wherein the generating the feedforward signal comprises: applying atransfer function to the control signal based on a negative relationshipbetween a voltage applied to a voltage controlled oscillator of thefirst phase-locked loop circuit and jitter of the intermediate outputsignal.
 22. The method of claim 19, wherein generating the intermediateoutput signal comprises: generating a phase difference signal based on adifference in phase between the input signal having the first frequencyand a phase-locked loop feedback signal; generating the control signalbased on the phase difference signal; generating the intermediate signalhaving a frequency adjusted based on the control signal; generating afirst phase signal and a second phase signal based on the intermediatesignal, both the first and second phase signals having a same frequencyas the intermediate signal, and the second phase signal phase-shiftedrelative to the first phase signal; generating a modulated phase controlsignal specifying a selected phase for each period of the modulatedphase control based on a frequency selection signal; generating aninterpolated signal for each period of the modulated phase controlsignal, the interpolated signal having the selected phase selected froma plurality of selectable phases in between phases of the first phasesignal and the second phase signal; and dividing a frequency of theinterpolated signal to generate the phase-locked loop feedback signal.23. The method of claim 19, wherein generating the output signalcomprises: dividing a frequency of the intermediate signal from thefirst phase-locked loop circuit to generate a frequency-divided inputsignal; generating a phase difference signal based on a difference inphase between the frequency-divided input signal and a phase-locked loopfeedback signal; generating a frequency control signal based on thephase difference signal and the feedforward signal; generating theoutput signal based on the frequency control signal; and dividing afrequency of the output signal to generate the phase-locked loopfeedback signal.
 24. The method of claim 23, further comprising:generating a quadrature signal having a phase shift relative to theoutput signal based on an oscillator feedback signal; determining aphase error between a detected difference in phases of the output signaland the quadrature signal; comparing the detected difference to adesired difference in phases; and generating the oscillator feedbacksignal to reduce the detected phase error.